Motor drive control

ABSTRACT

A control strategy for a multiple phase brushless motor (1) is disclosed in which the top (2, 3, 4) and the bottom (5, 6, 7) switching devices in two arms of the motor bridge are driven using complementary pulse width modulated waveforms, so that the top device (2, 3, 4) in one of the arms is in the ON state while the bottom device (5, 6, 7) is in the OFF state and vice versa. In an alternative arrangement, the control strategy described hereinbefore is used at high motor speeds while at low motor speeds a conventional modulation scheme is used in which a top switching device (2, 3, 4) in one arm of the bridge is in the ON state and a bottom device (5, 6, 7) in a different arm of the bridge is pulse width modulated. In another aspect, a method of calculating the position of the motor rotor is disclosed.

This is a United States national application corresponding to copendinginternational application PCT/GB97/00052, filed Jan. 9, 1997, whichdesignates the United States, the benefit of the filing date of which ishereby claimed under 35 U.S.C. § 120, which in turn claims the benefitof British application No. 9600549.1, filed Jan. 11, 1996, the benefitof the filing date of which is hereby claimed under 35 U.S.C. §119.

This invention relates to improvements in electrical motor controlstrategies, especially but not exclusively for use in an electricalpower steering system.

It is known in the art to provide an electrical power steering systemfor vehicles in which an electrical motor replaces the traditionalhydraulic assistance, so as to produce a compact, efficient steeringsystem. Steering assistance is provided by driving the motor in responseto a measure of the torque in the steering column. In such anapplication, it is essential to have good control of the outputcharacteristics of the motor, i.e. the motor torque.

A typical motor and drive circuitry of the kind to which this inventionrelates, and which may be used in a power steering system, comprisesthree elements; a multiple phase permanent magnet brushless motor inwhich the phases are connected in a star fashion, a multiple phasebridge, each arm of the bridge comprising a top switching means and abottom switching means; and sensing means for detecting rotor positionand/or motor current. The top and bottom switching means typically eachcomprise a single transistor or two or more transistors connected inparallel.

The control strategy controls the modulation technique used to drive thetransistors, thereby controlling the current in the motor windingstogether with knowledge of rotor position and hence controlling themotor torque.

Traditionally, such a motor system is driven by a dedicated integratedcircuit which performs the function of the control strategy, for examplethe Motorola MC33035 integrated circuit (IC). This provides all thefeatures necessary to implement a motor control strategy for a 3 or 4phase motor. The IC provides 3 outputs for driving the top transistors,and 3 outputs for driving the bottom transistors in each arm. For anygiven rotor position, the IC is used to enable only one top and onebottom transistor of the drive circuitry. An additional feature of thistype of motor control is that the enabled top and bottom transistors areon different arms of the bridge.

In the enabled state, the bottom transistor only is pulse widthmodulated, with the enabled top transistor being in the ON state for thewhole modulation period. Varying the pulse width modulation duty of thebottom transistor determines the motor voltage and can be used tocontrol motor current level and speed. The selection of the desired topand bottom transistor is performed under the IC's simple controlstrategy which interprets the data from a position sensor in conjunctionwith measurements of the current flowing in the motor windings. Thistype of modulation will be referred to as bottom only modulation,because only the bottom transistors are pulse width modulated.

A disadvantage of the conventional commutation including the bottomcontrol as described above is that in open loop voltage control mode, atransient drop in overall resultant motor current level (and hencetorque) occurs upon commutation from one phase to the next. In the powersteering application set forth, these transients are felt by the driverthrough the steering wheel. Other problems arise when the motor ischanging direction, due to poor current control using bottom-onlymodulation when the motor decelerates.

An aim of the present invention is to provide an improved motor controlstrategy which provides accurate control of motor torque irrespective ofrotor speed and position, to minimise motor torque ripple and tominimise acoustic noise from the motor.

According to a first aspect of the invention, in a control strategy forcontrolling the pulse width modulation of a multiple phase brushlessmotor having at least three phases in which each phase of the motor isconnected to an arm of a multi-phase bridge and each arm of the bridgecomprises a top switching device and a bottom switching device, saidswitching devices comprising at least one transistor,

said control strategy is characterised by comprising driving theswitching means with a first modulation mode in which complementarypulse width modulated inputs are applied to the top and bottom switchingdevices in two arms of the bridge so that in each of said two arms thetop switching device is in the ON state whilst the bottom device is onthe OFF state, and at the end of the pulse width modulation duty cyclethe state of the two switching devices is reversed,

selection of said pulse width modulation duty cycle and which switchingdevices are modulated being performed by a control algorithm in responseto the output of a sensing means.

In this control strategy, the pulse width modulated inputs applied tothe top and bottom transistors in a given arm of the bridge arecomplementary to one another, i.e. during a single modulation period,one transistor is ON whilst the other transistor is in the OFF state,and at the end of the pulse width modulation duty within the modulationperiod, the state of the two transistors is reversed. The averagevoltage applied to the motor is then determined by the ratio of the ONtimes of the two transistors. The state of the two transistors thereforeswaps during each modulation period.

This modulation technique will be known as complementary modulation, asthe transistors in each arm of the bridge are energised as thecomplement of each other. The direction of the applied motor voltage isreversed on each PWM cycle. Zero net motor voltage is applied at 50% PWMduty when forward voltage is applied for the same time as reversevoltage, and there is no discontinuity of the bridge operation uponreversal of net applied motor voltage. In this manner, controldifficulties experienced in the bottom only modulation technique areeliminated, and smooth reversals of the steering wheel can be achievedwhen used in an electric powered assisted steering (EPAS) system. Also,by using complementary mode switching the commutation transients aresymmetrical for both positive and negative half cycles of motor current,whereas this is not true in the case of lower or bottom-only control.

Preferably, an interlock delay can also be provided if this feature isnot provided in the driver IC for each transistor.

The addition of an interlock delay prevents "shoot-through".Shoot-through occurs when a top and bottom transistor in the same leg ofthe bridge are both turned on. MOS-FET and other transistors take afinite time to turn on and off so the interlock delay can be used todelay turning on of the bottom transistor after turning off the toptransistor (or vice versa).

Preferably, the sensing means may comprise a current sensor fordetecting the current flowing in the motor, and an angular positionsensor for detecting the rotor position of the motor rotor, the outputsof both sensors being employed by the control algorithm.

In this case, the motor current is controlled by reading the currentsensors, processing the measured values in accordance with the output ofthe position sensors to provide a `motor current` value, applying anappropriate feedback control algorithm that compares this `motorcurrent` to a demanded value in software, and adjusting the PWM duty inaccordance with the output from the control algorithm. The PWM duty isapplied to the appropriate transistors of the bridge according to theoutput of the angular position sensors.

Preferably, the motor may comprise a 3-phase motor and a 3-phase bridgewill then be used.

Hall effect devices can be used as the position sensors, and may bearranged in a manner that enables the output of the hall sensors to beexpressed as a 3-bit digital code. This digital code can then beemployed by the control strategy when determining which transistors willbe pulse width modulated.

In the complementary PWM mode, just two of the motor phases may beenergised at any given rotor position except for commutation transientswhen all 3 phases conduct for a short period, typically the same twophases as would be used in the bottom only modulation technique. In thisway, it is not necessary to use costly high resolution position sensors.

Whilst the above modulation scheme offers a considerable improvementover the bottom only modulation technique during motor reversals, adisadvantage of the complementary scheme is that it generates a higherripple current in the bridge filter capacitor.

Therefore, according to a second aspect of the invention, we provide acontrol strategy for controlling the pulse width modulation of amultiple phase brushless motor having three phases in which each phaseof the motor is connected to an arm of a multi-phase bridge and each armof the bridge comprises a top switching device and a bottom switchingdevice, said switching devices comprising at least one transistor,

characterised in that said control strategy selects from a firstmodulation mode and a second modulation mode:

whereby in said first modulation mode, the top switching device and thebottom switching device in at least one arm of the bridge are driven bycomplementary pulse width modulated inputs applied to the top and bottomswitching devices so that one of the said top or bottom switchingdevices is in the ON state whilst the other one of the top or bottomswitching device is in the OFF state, and at the end of the pulse widthmodulation duty cycle, the state of the two switching devices isreversed;

and whereby in said second modulation mode, a pulse width modulatedinput is applied to a bottom switching device in one of said arms ofsaid bridge whilst a top switching device in a different one of saidarms of said bridge is switched ON, selection of the pulse widthmodulation duty cycle and which switching devices are modulated beingperformed by a control algorithm.

Preferably, the control algorithm employs the information from thesensing means to determine the most suitable modulation technique.

Preferably, complementary modulation is used at low motor speeds to givegood control of motor reversals, and bottom only PWM is used at highmotor currents to minimise the ripple current in the bridge filtercapacitor. The control algorithm may switch motor drive from one mode tothe other at a motor current of approximately half the maximum motorcurrent.

The current flowing in ground return from the bridge and in each of themotor phases and/or the individual phase legs can be sensed by means ofthe potential drop across a series resistor in these paths.

A third aspect of the invention is the implementation of a softcommutation mode or a hard commutation mode when either complementarymodulation or the bottom only modulation is selected by the controlalgorithm.

In a soft commutation mode, the rate of the decrease of the terminatingphase current is decreased to match the rate of increase of thecommencing phase current. This offers the advantage of maintaining thecurrent in the third phase at a constant level without the undesirablecurrent transient that otherwise occurs due to inductive effects. Thereduced rate of change at the commutation point reduces the magnitude ofthe induced EMF's in the motor phases and also reduces acoustic noise.

Preferably, the duration of the soft commutation is controlled by thecontrol algorithm in response to motor current and rotor position so asto optimise performance at low rotor speed and to minimise the drop inmotor torque at high rotor speed.

Alternatively, hard commutation may be employed. In this case, the rateof increase of the commencing phase current is increased to match therate of decrease of the terminating phase current to maintain a constantcurrent in the third phase, and again overcoming the undesirablemomentary drop in current (and hence torque) that otherwise occurs. Theduration of such a hard commutation may be controlled by the controlalgorithm or in software.

A second aim of the present invention is to provide a means fordetermining the absolute position of the rotor of the motor, enablingfurther improvements in detection and control of the commutation event.

According to a fourth aspect of the invention, a method of calculatingthe position at a moment in time of a motor which is connected to anoutput shaft through an intermediate means comprises the steps of:

obtaining a first measurement of actual rotor position at a firstinstance in time using sensing means provided at the motor;

calculating speed of rotation of said output shaft using a sensing meansprovided on said output shaft;

calculating an offset indicative of the displacement of the rotor of themotor between the first instance of time and said moment in time basedupon the speed of rotation of the output shaft, and;

modifying the first measurement of rotor position obtained at the firstinstance by adding the offset to produce an output indicative ofabsolute motor rotor position at said moment in time.

Preferably the motor position sensing means comprises a high precisionabsolute motor position sensor.

Alternatively the motor position sensing means comprises a combinationof position measurements from sensing means on the output shaft with theoutput of position sensors on the motor.

Alternatively the motor position sensing means comprises a combinationof absolute position sensor on the output shaft with an incrementalposition sensor on the rotor of the motor.

Preferably, the motor position sensor may comprise Hall effect sensors,which may produce a 3-bit digital code. In the case of an electricalpower steering system, for example, the intermediary means may comprisea clutch and/or a gearbox, and the motor may drive the output shaftthrough a worm and worm wheel.

The position of the output shaft, which in the case of an electricalpower steering system would be the steering column, can be obtained froma dedicated position sensor or from a suitable output from a torquesensor, such as the Lucas Linear Array Torque sensor.

The rotor position sensors, i.e. Hall effect sensors, provide anabsolute indication of motor electrical angle. Whenever a Hall sensorchanges state, an offset is set to zero. On every periodic sample, theoutput shaft velocity can be determined from the output shaft positionsensor. This may be done by comparing the last two shaft positionmeasurements. The gearbox ratio may be multiplied by the shaft velocityand added to this offset. The Hall effect state plus the offset enableshigh resolution absolute position information to be determined.

In this manner, by resetting the offset on every Hall sensor transitionthe effects of backlash and torsional wind-up is minimised, and so thatthe position information on the output shaft can be used to indicate theposition of the motor rotor.

According to a further aspect of the present invention, absolute angularposition information can be used to optimise the motor commutationposition under different motor operating conditions.

The commutation position may be varied by advancing or retarding theposition at which a commutation event occurs, in response to theabsolute motor position readings. This minimises the torque step thatcan occur at the commutation point, due to changes in the magneticeffects in the motor as current levels increase. The commutation pointcan also be varied as a function of the motor speed to provide highertorque at higher motor speeds, thus improving the steady statetorque/speed envelope of the motor for a given battery voltage.

The demanded current may also be controlled in order to minimisepredictable on-load motor torque ripple by adjusting the current inresponse to motor current, motor velocity and absolute positioninformation.

Some embodiments of our invention are illustrated in the accompanyingdrawings in which:

FIGS. 1a and 1b show a 3-phase brushless DC motor and associated powerdrive circuitry which comprises a 3-phase bridge;

FIG. 2 shows the bottom only modulation technique used and the currentstate of each transistor for a given Hall sensor and code;

FIG. 3 illustrates the complementary PWM strategy according to a firstaspect of the invention;

FIG. 4 illustrates the two PWM1 and PWM2 signals;

FIG. 5 illustrates a comprehensive control strategy which employscomplementary and bottom only modulation according to the second aspectof the present invention;

FIG. 6 shows the Hall sensor code changes which result in topcommutation;

FIG. 7 shows the Hall sensor code changes which results in bottomcommutation;

FIG. 8 shows an example of the bottom soft commutation strategy;

FIG. 9 shows an example of a control routine for implementing bottomsoft commutation;

FIG. 10 shows an example of the top soft commutation strategy;

FIG. 11 shows an example of a software control routine for implementingtop soft commutation strategy;

FIG. 12 illustrates the relationship between Hall effect sensors and themotor armature such that rotor position information can be obtained;

FIG. 13 illustrates schematically an electrical power steering systemwhich embodies the control strategy of FIG. 5;

FIGS. 14(a) and (b) illustrates the use of current shaping by applying aboost voltage to minimise position dependent ripple; and

FIG. 15 shows the relationship between the transient dip in current inthe positive A and negative B cycles for (a) the complementary and (b)bottom only modulation schemes.

It is well known that the instantaneous torque in a permanent magnetexcited electric motor can be controlled so that it is substantiallyproportional to the instantaneous motor current in the excited phases.Hence, controlling the motor current also controls the motor torque.Since it is cheaper to measure motor current than to directly measuremotor torque, a current control system is implemented.

A specific embodiment of a motor and power drive circuitry is shown inFIG. 1. The motor (1) shown is a three-phase brushless DC motor, thephases of the motor being connected together in a star configuration.The drive circuitry comprises a three phase bridge. Each arm of thebridge further comprises a pair of transistors connected in seriesbetween a supply rail and a ground, with the motor windings being tappedoff from between the two transistors. MOS-FET type transistors are used.The transistors in each arm are referred to in this text as toptransistors (2,3,4) and bottom transistors (5,6,7) respectively.

This type of motor and drive circuit is often controlled using adedicated IC such as Motorola MC 33035 that switches one top fieldeffect transistor (FET) fully on, and applies a pulse-width modulated(PWM) control signal to one bottom FET. The particular FETs aredetermined by the IC which decodes information from an angular positionsensor in conjunction with current information.

In one example the angular position of the motor rotor is sensed throughuse of magnetic effect sensors placed so as to detect the passing of anoverhang section of the rotor magnets wherein the sensor is arranged tohave a magnetic shield that provides a magnetic path for the sensorcircuit itself whilst shielding the sensor from the effects from anyexternal magnetic fields.

In operation the sensing element switches each time a magnetic circuitis made or broken which occurs as a pole of the motor or rotor magnet 27passes a sensing element. This switching signal is passed viaconnections, not shown, to the motor control circuit.

The rotor carries a back iron sleeve 26 having magnets 27 secured aroundits periphery covered by a rotor sleeve 28. A clearance or air gap 29 isprovided to allow the rotor to move angularly with respect to the sensorassembly 30 and motor stator (not shown).

The sensor assembly shown in FIG. 12 is a three element devicesurrounding an arc or outer section of the rotor. The size and number ofelements 31 within the sensor assembly may vary between 1 and any numbercovering the entire circumference of the rotor, depending upon theconfiguration i.e. the number of rotor pole pieces of the motor.

The sensor assembly comprises a non-magnetic sensor carrier 32 in whichare placed the magnetic sensing elements 31. This is surrounded by amagnetic sensor back iron 33 which acts to enhance the magnetic fieldstrength around the sensing elements to aid switching and eliminatemagnetic interference, and importantly to provide the sensor assemblywith its mechanical robustness, as typically the materials from which asuitable non-magnetic sensor carrier would be made are low in mechanicalstrength.

The output signal from the Hall-effect sensing means can be adapted toprovide a 3-bit digital code. The need for the 3-bit code is optional,and used in the preferred embodiments disclosed in this application byway of example only. At least one or more sensors providing one or morebits of position information could be used subject to the design andperformance and fault tolerance constraints imposed on the actualworking application.

Information about the motor current is obtained by measuring the currentflowing in ground return from the bridge and/or in each of the 3 motorphases.

In FIG. 1b, only a single sensing resistor 8 is shown. Measuring thecurrent flowing in the resistor can be performed by way of measuring thepotential drop across the resistor and applying Ohm's Law. This singleresistor provides information on the current flowing in ground returnfrom the motor.

Alternatively, a resistor 8a,8b,8c can be provided for each arm of thebridge as shown in FIG. 1b. This allows the current in each phase to bemeasured, i.e. phase leg sensing.

Both the current and position information can be used by the controlstrategy as below.

The motor current is controlled by reading the current sensors,processing the measured values in accordance with the 3-bit digital codefrom the Hall-effect sensors to provide a motor current value, applyingan appropriate feedback control algorithm that compares this "motorcurrent" to a demanded value in software, and adjusting the PWM duty inaccordance with the output from the control algorithm. The PWM duty isapplied to the appropriate transistors of the 3-phase bridge accordingto the 3-bit digital code from the Hall effect sensors. This method ofmotor control is advantageous as it requires fewer components thanelectronic hardware, assuming that the microprocessor is alreadypresent.

One form of motor modulation technique which can be used with the abovecontrol strategy is known as bottom-only PWM modulation, as in all motorcontrol circumstances only the bottom transistors 5,6,7 are pulse widthmodulated, whilst the top transistors 2,3,4 are either in the ON stateor the OFF state, as determined by the control strategy. FIG. 2 shows abottom-only modulation scheme showing the state of the transistors ineach arm of the bridge for a given Hall effect switch code. It can beseen that in this modulation technique, only a single modulationchannel, PWM1, is required.

A disadvantage of the above control strategy, in which conventionalcommutation is employed, is that there is a transient drop in the motorcurrent (and hence torque) in the uncommutated phase upon commutation ofthe other two phases. This originates from attempting to enforce fastchanges of current in the inductive windings of the motor. Thistransient drop in motor torque, which occurs upon commutation, can befelt and heard at the steering wheel when the motor is operating in anEPAS system.

A further disadvantage of the bottom-only PWM technique is that itprovides poor control of current when the motor is reversed. The mostimportant reversal is one in which the motor current sign is changed sothat instead of putting mechanical power into the steering system, themotor is taking mechanical power out of the steering system (that is,braking the movement of the output shaft). Under such reversals, it isdifficult to accurately control the motor current with bottom-only PWM.

This is because the bridge has a non-linear response under such areversal. When the motor is braking, regenerative currents are generatedin the motor windings. With bottom-only switching, these regenerativecurrents can only flow when the bridge is reversed (i.e. when the toptransistors are switched). Thus when the bridge is reversed, theregenerative current starts to flow very suddenly causing a sharp changein motor current (and hence motor torque). Such a sharp change in torqueis undesirable.

In an EPAS system it is essential to have good current control undermotor reversal and the motor current is often reversed with the motormoving (and hence braking).

FIG. 3 provides details of an example of an alternative PWM mode knownas complementary PWM mode which forms a first aspect of this invention.It is known as complementary PWM because the transistors are pulse widthmodulated as the complement of each other using complementary PWMsignals. In this modulation scheme, the direction of the applied motorvoltage is reversed every PWM cycle. It can be seen, therefore, that thetransistors are energised as a complement to each other, and both thetop and bottom transistors can be modulated. It can be seen that two PWMchannels, PWM1 and PWM2 are needed.

The two PWM channels have detail characterising features as shown inFIG. 4. PWM-PERIOD 50 is the total time of one modulation cycle.PWM1-HIGH-TIME 51 is the duty time that PWM1 is high (i.e. transistorturned `ON`) and PWM2-HIGH-TIME 52 is the time that PWM2 is high (i.e.transistor turned `ON`). In normal complimentary operation, the twochannels are complimentary and only one transistor is on at a time, thetransistors swapping over every during each modulation period. AnINTERLOCK delay 53 is also provided to prevent shoot through.

In operation, zero net motor voltage is applied at 50% PWM duty whenforward voltage is applied for the same time as reverse voltage. Thus,in this mode there is no discontinuity of bridge operation upon reversalof net applied motor voltage and hence no torque transient is generated.This results in smooth reversals of the steering wheel.

A similar PWM, mode is commonly employed in AC brushless drives, where a`flux` vector is established by energising all three motor phases to adifferent degree. However, this requires a high resolution positionsensor which is costly. In the complementary PWM technique, only twophases are energised at a given rotor position, and so low resolutionposition information, such as the 3-bit code generated by the Hallsensors, can be employed.

The complementary PWM technique offers the advantage that the transientswhich occur during bottom only modulation are eliminated. However, adisadvantage of the complementary modulation technique is that itgenerates higher ripple current in the drive circuit than the bottomonly modulation.

In view of the disadvantages and advantages between the two PWM modes,the control strategy shown in FIG. 5 employs a combination of bottomonly modulation and complementary modulation. Complementary modulationis selected for currents of low value either side of zero (e.g. ±10amps) to give smooth control of motor reversals. Bottom only modulationis then switched in by the control strategy for higher currents tominimise current ripple in the drive stage. Switching between the two isperformed by the control algorithm in connection with the currentsensing means. This dual modulation control strategy forms the secondaspect of the invention.

FIG. 13 illustrates an electrical power steering system whichincorporates a motor control strategy of the kind illustrated in FIG. 5.An electronic control unit 34 is adapted to receive signals from anignition switch 35, vehicle speed signalling means 36 diagnostic means37, CAN interface means 38, and torque sensor electronic means 39. Theelectronic control unit 34 operates on the various signals and emits anenergising current to control the power assistance applied to a steeringmechanism 40.

As illustrated a steering wheel 41 controls the operation of a steeringlinkage 42 through a column shaft 43. The torque applied to the columnshaft 43 is augmented by an electric actuator 44 under the control ofthe energising current from the electronic control unit 34.

FIG. 5 illustrates various logical means, in software terms, that wouldbe embodied within the controller shown in FIG. 13. In this controlstrategy, both complementary 9 and bottom only 10 modulation areemployed.

Complementary modulation is used at low motor currents, typically up tohalf the maximum motor current. If the output from the current sensingmeans shows that the current exceeds a pre-set value, bottom onlymodulation is employed. The control strategy will switch modulation backto complementary modulation if the current falls back below this pre-setvalue.

The control strategy when bottom only modulation is selected, employseither bottom soft commutation 11 or top soft commutation 12. Selectionof the correct commutation strategy is performed in response to whethera bottom commutation or a top commutation occurs. On completion ofeither the top or bottom commutation event, the control strategy returnsto normal bottom only modulation 10.

The control strategy can also switch directly back to complementarymodulation from either the bottom soft commutation 11 or top softcommutation 12.

In bottom-only modulation, two types of commutation event can occur. Atop commutation occurs when the Hall sensors change state resulting inone top FET turning off and a different top FET turning on. The Hallsensor code changes when this form of the commutation occur as shown inFIG. 6.

A bottom commutation occurs when one bottom FET is turning off and adifferent bottom FET is turning on. The Hall sensor code changes whenthis form of commutation occurs are shown in FIG. 7.

In order to understand the purpose of the two forms of soft commutation,the behaviour of the motor under a normal commutation must beappreciated. The commutation event consists of terminating the currentflow in one motor phase, commencing an equivalent current flow in asecond motor phase and maintaining a nominally constant current in athird motor phase. Since the three motor phases are connected in a starconfiguration to a star point, the sum of the currents flowing in thethree phases must remain at zero. However, the rate of increase of thecommencing phase current is less than the rate of decrease of theterminating phase current, and so the third (normally constant) phasecurrent drops momentarily, whilst maintaining a value that makes the sumof the three phase currents equal to zero. This drop corresponds to atransient drop in motor torque.

One solution is to increase the rate of increase of the commencing phasecurrent so as to match the rate of decrease of the decreasing phasecurrent to maintain a constant third phase current and hence a constantmotor torque. This is referred to as hard commutation.

This hard commutation effectively uses a `voltage boost` feature wherebythe PWM duty is increased for a short period while commutationtransients are present. In bottom only mode the current transient dip inthe uncommutated phase are not symmetrical for both positive andnegative directions of current in the phase. This is shown in FIG. 15.Therefore a different level of voltage boost is required to minimise thecurrent dip depending on the direction of current. In complementary modethe dip in the uncommutated phase in the same for both directions ofcurrent flow and therefore only one level of voltage boost is needed fora given current level in either direction.

Another solution is to seek to decrease the rate of decrease of theterminating phase current to match the rate of increase of thecommencing phase current. When the two are matched, the resultingcurrent in the third phase remains constant during the commutation eventand hence the motor torque also remains constant. This solution isreferred to as soft commutation.

FIGS. 8 and 10 show the control strategy employed in order to effectbottom soft and top soft commutation in response to the Hall sensoroutput code changes during bottom only modulation.

In a refinement of the bottom soft and bottom hard commutationtechniques, the duration of the commutation, i.e. the number of PWMcycles over which the commutation is effected, is adjusted undersoftware control. This allows the commutation function to be optimisedat low rotor speeds and minimises the drop in motor torque at high rotorspeeds. This is illustrated in FIGS. 8 to 11 for the case of softcommutation.

The soft and hard commutation techniques can be implemented by varyingthe pulse width modulation period PWM2-HIGH-TIME. In the case of bottomsoft and top soft commutation, the value of PWM2-HIGH-TIME can be variedaccording to the control strategy in response to motor torque andvelocity.

An example of the implementation of bottom soft modulation in thismanner is shown in FIGS. 8 and 9. On entry to the bottom soft modulationstate, PWM2-HIGH-TIME is initialised to the same value asPWM1-HIGH-TIME. It is then progressively reduced to zero over apredetermined number of PWM cycles. The number of PWM cycles is thusdetermined by the control unit according to the motor operatingconditions.

In the example provided in FIG. 9, the control strategy bottom softcommutation sub routine determines the number of PWM cycles over whichPWM2-HIGH-TIME is reduced as follows. After initialising PWM2-HIGH-TIME13, the control strategy checks to see if the next PWM cycle has beenreached 14. When the next cycle is reached, the control strategy movesonto the next step which is to determine whether or not a commutationhas occurred 15. If a commutation has occurred, the bottom softcommutation routine is exited 18. If commutation has not occurred,however, PWM2-HIGH-TIME is reduced 16 by an amount which is dependentupon the motor torque and the motor velocity. If, after this reduction,PWM2-HIGH-TIME is less than or equal to zero 17, the bottom softcommutation mode is exited. If PWM2-HIGH-TIME is still greater thanzero, the routine returns to check whether the next PWM cycle has beenreached 14. This is repeated until PWM2-HIGH-TIME is equal to or lessthan zero and bottom soft commutation is exited.

The bottom soft commutation state is exited normally when PWM2-HIGH-TIMEreaches zero, and the rising edge of PWM1 and the falling edge of PWM2remains synchronised at all times.

A similar modulation technique can be implemented for the top softcommutation, shown in FIGS. 10 and 11. In this case, PWM2x-HIGH-TIME isinitialised to the same as PWM-PERIOD on entry to this state. It is thenprogressively reduced to zero over a predetermined number of PWM cycles.As in the case of bottom soft commutation, the number of cycles ispredetermined and exit from this state occurs when PWM2-HIGH-TIMEreaches zero. The rising edge of PWM1 and the falling edge of PWM2remain synchronised.

FIG. 11 provides an example of a top soft commutation subroutine,expressed in terms of the logical means in software terms that could beembedded within a control strategy in order to implement top softcommutation. After first setting the PWM mode to top soft commutation19, PWM2-HIGH-TIME is initialised to the same value as PWM1-HIGH-TIME20. The control routine then checks whether the next PWM cycle has beenreached 21. If not, the routine remains in this state until the nextcycle is reached, whereupon the control routine enquires whether acommutation has occurred 22. If a commutation has occurred, top softcommutation is exited 25. If no commutation has occurred, the rate ofPWM2-HIGH-TIME is reduced 23 by an amount dependent upon the motortorque and the motor velocity. The control routine then checks 24 to seeif the new value of PWM2-HIGH-TIME is equal to or less than zero. Ifyes, top soft commutation is exited, but if the new value is greaterthan zero, the routine returns to the state where it waits for the nextPWM cycle to be reached 21.

In another refinement to the motor control strategy, high resolutionabsolute motor rotor position measurements can be employed to enable thecontrol system algorithm to optimise the rotor position at which themotor current is switched from one set of phase windings to another,i.e. the position of each commutation.

Typically, the position of the rotor at which commutation occurs isdetermined by the position of the Hall sensors, commutation beinginitiated by a change in the Hall sensor output code. Because the Hallsensors are attached to the stator, these commutation positions arefixed. Under certain conditions the motor performance can be improved byphysically moving the commutation positions with respect to the stator,which can help to reduce or eliminate torque discontinuities which ariseat commutation. As the motor current rises, the electromagnetic field ofthe stator distorts the magnetic field of the rotor. This distortioneffectively shifts the working angle of the phase. The distortion can becompensated by advancing the commutation point in the direction of themotor torque. The amount of advance needed depends on the magnitude ofthe motor current and when the motor current is negative the advance isalso negative (i.e. a retardation).

FIGS. 14(a) and (b) shows how motor current variations can be minimisedthrough the application of a current shaping technique. FIG. 14(a) showsthe motor torque without shaping. In FIG. 14(b), a voltage boost signalis applied to the motor, this signal being the mirror image, or inverse,of the motor torque ripple which would occur without smoothing (i.e.torque shown in FIG. 14(a)). The resultant motor current in this case isnominally free of ripple because it is the sum of the position dependentcurrent and the applied boost signal, as shown in FIG. 14(b).

The commutation point can also be varied as a function of motor speed toprovide higher torque at high motor speeds and thus improving the steadystate torque/speed envelope of the motor for a given battery voltage.

Using high resolution absolute motor position data, the commutationpoint can be changed under software control. This allows the optimumcommutation position to be selected for each motor operating condition.

High resolution position information can be provided by using the Halleffect switches in conjunction with an angular position measurementobtained from a torque sensor provided in the EPAS system to providesteering torque data. The operation of this absolute positionmeasurement is described in the following paragraphs.

The angular position of the motor rotor is measured by the Hall-effectsensors. The position information has a resolution of 360°/(3×number ofmotor poles) e.g. 20 degrees in a 6-pole motor. Angular position dataprovided by the torque sensor provides information about the angularposition of the worm wheel on the steering shaft. When the motor clutchis engaged, the motor is physically linked to the worm wheel through aworm and the clutch gearbox. Hence the angular position information fromthe torque sensor provides angular position information about the motorrotor. The angular position of the worm is known with respect to anarbitrary start position by resetting an offset to zero every time aHall sensor changes state, and the torque sensor provides positioninformation relative to this arbitrary position, so absolute position ofthe motor can be determined.

By referencing the measured angular position of the motor to the fixedangular position of a particular Hall-effect sensor transition (i.e.commutation position), it is possible to calculate an absolute motorposition with high accuracy. As long as the angular position from thetorque sensor is referenced to the Hall-effect sensors on everycommutation, it is possible to compensate for the effect of backlash inthe gear box and for torsional wind-up in the worm and wheel underconditions of high load. A suitable torque sensor for providing theangular position information required is the Lucas linear array torquesensor or the eight channel sensor.

In another embodiment it is known that certain components of the motortorque ripple are predictable, being related to the angular position ofthe rotor. The high resolution absolute motor position informationdescribed above can be used to determine a correction to the demandedmotor current in order to compensate for this ripple component. In thismanner the ripple can be mapped out.

Additionally, by observing when a commutation event is signalled by theHall effect sensors, a velocity value obtained from the last 2 highresolution absolute motor position readings can be used to calculate adelay time. For example, if a 2° advance angle is required then thevelocity measurement can be used to determine how long it will take totravel 2°. This time delay can then be clocked by the control strategyand used to trigger the commutation event after the elapsed time. Thisproves most effective when the commutation point is close to the Halleffect signal.

It will be understood from the above description that in one of itsaspects the present invention relates to an improved motor controlstrategy for an electric motor suitable for use in an EPAS system, inwhich two modulation techniques (drive modes) are employed depending onmotor conditions. In one mode, bottom-only modulation occurs, whilst inthe other mode, complementary modulation occurs, the choice of drivemode being selected by a control algorithm. Several enhancements arealso described to improve motor torque output. In this manner,considerable improvements over a basic control strategy are effected,resulting in an improved torque output from the motor.

The improved motor control strategy is described in the context of apower steering application. It is, however, to be understood that thecontrol strategy is suitable for use in any application in which it isdesirable to provide an improved torque output from an electric motor.

We claim:
 1. A control strategy for controlling the pulse widthmodulation of a multiple phase brushless motor having at least threephases in which each phase of the motor is connected to an arm of amulti-phase bridge and each arm of said bridge comprises a top switchingdevice and a bottom switching device, said switching devices comprisingat least one transistor, and a sensing means having an output, whereinsaid control strategy comprises driving said switching devices with afirst modulation mode in which complementary pulse width modulatedinputs are applied to said top and bottom switching devices in two armsof said bridge so that in each of said two arms said top switchingdevice is in the ON state whilst said bottom device is on the OFF state,and at an end of said pulse width modulation duty cycle the state ofsaid two switching devices is reversed, selection of said pulse widthmodulation duty cycle and which of said switching devices is modulatedbeing performed by a control algorithm in response to said output ofsaid sensing means.
 2. A control strategy for a motor according to claim1, in which said sensing means comprises at least one current sensor fordetecting the current flowing in said motor and an angular positionsensor for detecting the rotor position of said motor rotor.
 3. Acontrol strategy according to claim 1, in which the motor is used in anelectric powered assisted steering EPAS system.
 4. A control strategyfor controlling the pulse width modulation of a multiple phase brushlessmotor having three phases in which each phase of the motor is connectedto an arm of a multi-phase bridge and each arm of said bridge comprisesa top switching device, and a bottom switching device, said switchingdevices, comprising at least one transistor, wherein said controlstrategy selects from a first modulation mode and a second modulationmode:whereby in said first modulation mode, said top switching deviceand said bottom switching device in at least one arm of said bridge aredriven by complementary pulse width modulated inputs applied to said topand bottom switching devices so that one of the said top and bottomswitching devices is in the ON state whilst the other one of the top orbottom switching device is in the OFF state, and at the end of a saidpulse width modulation duty cycle, the state of the said two switchingdevices is reversed; and whereby in said second modulation mode, a pulsewidth modulated input is applied to said bottom switching device in oneof said arms of said bridge whilst said top switching device in adifferent one of said arms of said bridge is switched ON, selection ofsaid pulse width modulation duty cycle and which of said switchingdevices are modulated being performed by a control algorithm.
 5. Acontrol strategy according to claim 4, in which complementary pulsewidth modulation inputs are applied to said top and bottom switchingdevices in two arms of said bridge when said first mode is selected,said two arms being the same two arms as said arms containing saidbottom pulse width modulated switching device and said top ON switchingdevice when said second modulation mode is selected.
 6. A controlstrategy for a motor according to claim 4, in which said controlalgorithm switches motor drive from one of said modulation modes to theother at a motor current of approximately half maximum motor current. 7.A control strategy for a motor according to claim 4, in which one of ahard commutation mode and a soft commutation mode is employed.
 8. Acontrol strategy for a motor according to claim 7 in which softcommutation is implemented by decreasing the rate of decrease of theterminating phase current to match the rate of increase of thecommencing phase current.
 9. A control strategy for a motor according toclaim 8 in which the duration of the soft commutation is determined bymotor current and rotor position.
 10. A control strategy for a motoraccording to claim 9 in which hard commutation is employed by increasingthe rate of increase of the commencing phase current to match the rateof decrease of the terminating phase current.
 11. A control strategy fora motor according to claim 10 in which the duration of the hardcommutation is controlled by software.
 12. A control strategy for amotor according to claim 4, in which the commutation position is variedby software by advancing or retarding the position at which acommutation event occurs in response to motor position readings.